1. Field of the Invention
This invention relates generally to a technique for determining the power linearity of a detector and, more particularly, to a technique for determining the power output linearity for a detector associated with a microwave radiometer.
2. Discussion of the Related Art
Certain detectors, such as microwave radiometers, receive electromagnetic noise radiation from a remote scene, and convert this radiation to a DC voltage representation so as to generate, for example, a temperature gradient image of the scene. Microwave radiometers receive thermal noise radiation from the scene and generate a DC voltage equivalent of the noise at microwave radio frequencies (RF). This principal is possible because, as is well understood, RF thermal noise being radiated from a body is proportional to the temperature of the body. Because the amount of radiation a body will generate is proportional to its temperature, it is necessary that the voltage output of the detector be substantially linear in order to give an accurate characterization of the temperature of the body. A microwave radiometer, therefore, relies on the power linearity of its microwave receivers to accurately measure the remote microwave thermal noise sources. In some applications, rigid accuracy is not necessary, and thus the linearity of the output of the receiver can be more relaxed. However, in other applications, the accuracy is very important, and thus, the output linearity of the receiver is critical.
One application in which receiver output linearity is crucial for generating accurate output data is realized in a tropical rainfall measuring mission (TRMM) microwave imager (TMI). The TMI incorporates a microwave radiometer which relies on the power linearity of its microwave receivers to produce noise temperature scans of the earth at a series of frequency and polarization channels. FIG. 1 shows a block schematic representation of one channel of a receiver 10 associated with a microwave radiometer used in a TMI. The receiver 10 is based on a satellite (not shown) orbiting the earth 12 in a geosynchronous orbit above a predetermined area of the earth 12. Electromagnetic thermal noise radiation 14 from the earth 12 in the microwave frequency range is received by the receiver 10. An earth reflector 16 mounted on the satellite receives the electromagnetic thermal noise radiation 14 and reflects this radiation onto an antenna feed device 18 as indicated by an arrow designated T. The earth reflector 16 is continually rotating about an axis such that a scan line across the earth 12 is generated. As the feed device 18 is rotated, thermal radiation T.sub.c from a cold source 20 and thermal radiation T.sub.h from a hot source 22 are also intermittently applied to the feed device 18. The temperature values T.sub.c and T.sub.h provide known cold and hot thermal noise signals to the receiver 10 for calibration purposes in order to compensate for certain things such as drift in the receiver 10, as is well understood in the art.
The electromagnetic radiation 14 from the earth 12, the cold source 20 and the hot source 22 are applied by the feed device 18 to a low noise amplifier (LNA) 24 in order to provide an amplified receiver thermal noise input signal. The amplified noise signal from the amplifier 24 is then applied to a bandpass filter 26 in order to limit the noise signal to a particular frequency range representative of the particular channel so as to establish a noise or convolution bandwidth. A diode detector 28 receives the bandwidth noise signal from the bandpass filter 26 and generates a DC voltage value indicative of the noise signal. The voltage value is applied to a video amplifier 30 in order to generate an output voltage V that varies with the microwave thermal power noise signal applied to the feed device 18.
FIG. 2 shows a graph of the relationship between the receiver output voltage V at the output of the video amplifier 30 on the vertical axis and the RF thermal noise input signal on the horizontal axis. The point labeled A represents the power input and associated voltage output point of the cold source 20, and the point labeled B represents the power input and associated voltage output point of the hot source 22, where V.sub.h is the hot load voltage output of the receiver 10 when the feed device 18 is directed towards the hot source 22 and V.sub.c is the cold load voltage output of the receiver 10 when the feed device 18 is directed towards the cold source 20. The actual relationship between the output voltage and the input noise signal RF power point for a particular input as the reflector 16 scans the earth 12 will be found along the solid curved line between the points A and B.
The power linearity requirement falls principally on the detector diode 28 used to convert the RF noise signal to a DC voltage. Different diode detectors 28 will have different output curves between points A and B. Therefore, it is not known what the actual output voltage would be for a particular noise input signal received from the scan of the earth 12. Because the output voltage of the detector 28 is nearly linear, the output voltage V for a particular RF noise signal received by the feed device 18 is generally represented by an interpolation algorithm to calculate V along a linear line between the points A and B, as shown by a dotted line, based on the thermal noise radiation input signal. Therefore, an error .delta.V is introduced as a difference between the linear interpolated voltage value and the actual voltage value from the diode detector 28. Consequently, it is sometimes necessary to determine the linearity of the output of the receiver 10 in order to assess whether this linearity is adequate enough for the desired application.
The thermal noise power as applied to the feed device 18 is represented by: EQU P=kB[T+(F-1)T.sub.o ], (1)
where,
T is the noise temperature of the body being detected; PA1 F is the receiver noise; PA1 B is the receiver convolution (noise) bandwidth; PA1 k is Boltzman's constant; and PA1 T.sub.o is 290.degree. K. PA1 T.sub.c is the temperature of the cold source 20. PA1 .delta.P=kB.delta.T; and PA1 P.sub.h is the input power at T.sub.h.
By measuring the output voltage V when the antenna feed device 18 is pointed towards the cold source 20, the hot source 22 and the earth 12, T can be determined as an estimate of T by using the linear interpolation formula: ##EQU1## where, T.sub.h is the temperature of the hot source 22; and
Typically, the linear interpolation must determine T to a fraction of a degree K out of a system noise temperature of 1000K or more. If .delta.T is defined as the worst case difference between the interpolated temperature T and the actual temperature T, it can be shown from FIG. 2 and equation (1) that: EQU .delta.T/T.sub.h .congruent.F.delta.P/P.sub.h, (3)
where,
This power linearity .delta.P is required over a power range given by: EQU R=P.sub.h /P.sub.c .congruent.F/(F-1). (4)
Table I shows the power linearity requirements for five channels of a known TMI. In Table I, two quantities, .delta.P.sub.h and 1+.delta.P/P.sub.h, are listed to express the power linearity requirements. The first quantity .delta.P/P.sub.h is useful as a measure of the nonlinear power distortion of the receiver 10. The second quantity 1+.delta.P/P.sub.h is useful as a measure of the power linearity required for the temperature interpolation. Note that a 1+.delta.P/P.sub.h of about 0.001 dB is needed to meet .delta.T requirements.
TABLE I ______________________________________ Detector Channel Temperature Noise Power Dis- Power Frequency Nonlinearity FIG. Range tortion Linearity f.sub.o .delta.T F R .delta.P/P.sub.h 1 + GHz K dB dB dB .delta.P/P.sub.h ______________________________________ dB 10.65 0.275 4.03 2.19 -34.26 0.0016 19.35 0.390 7.99 0.75 -36.70 0.0009 21.3 0.445 6.61 1.07 -34.75 0.0015 37 0.385 9.55 0.51 -38.32 0.0006 85.5 0.560 10.55 0.40 -37.69 0.0007 ______________________________________
Concerns about the power linearity of detectors have led to extensive use of double sideband (DSB) homodyne and single sideband (SSB) heterodyne receivers in previous radiometer designs. In both the DSB homodyne and SSB heterodyne receivers, a local oscillator (LO) is utilized to down convert the noise input signal to an intermediate frequency (IF) before the signal is detected. The principal advantage of these receivers is that the same detector can be utilized for all frequency bands. However, the receiver 10 of FIG. 1 is much simpler than either of the DSB homodyne and SSB heterodyne receiver designs, and thus offers significant cost and complexity advantages. The principal disadvantage, however, of the receiver 10 is that the detector 28 must operate at the input RF frequency, so a different diode detector 28 must be used for each frequency band.
Because of the simplicity of the receiver 10, it is desirable to be utilized as much as possible in radiometer receivers. This principally depends on finding suitable linear RF detectors at each of the desired frequency ranges and on developing simple low-cost methods for verifying the power linearity of RF detectors. What is needed then is a low cost and simple technique for verifying the power linearity of RF detectors. It is therefore an object of the present invention to provide such a technique.